Michael J. Chudobiak

Avtech Electrosystems Ltd., PO Box 5120, Stn. F, Ottawa, Ontario, Canada K2C 3H4

Abstract

A dc-coupled non-linear pulse amplifier circuit is presented. The circuit presented can produce 40 V peak-to-peak pulses with 3 ns rise and fall times. This speed is obtained by using Class D transistor amplifier stages. This circuit is shown to be useful for measuring the reverse recovery transients of fast switching diodes such as the 1N4148, and fast recovery power rectifiers.

I. Introduction

The fastest op amps available today, such as the Comlinear¹ CLC203 are capable of generating pulses with 20 V peak-to-peak amplitudes and 4 ns rise times. Sub-nanosecond switching speeds can be obtained for amplitudes of up to 100 V or higher by using step recovery diode pulse sharpening circuits². However, in many cases this method is needlessly expensive and too fast, since very careful physical circuit construction is required to avoid inductive ringing and electromagnetic interference. Between these two circuit approaches there are relatively few circuits that will allow the generation of pulses of several tens of volts amplitude in 50 loads, with rise times of a few nanoseconds.

This paper presents a circuit that uses Class D transistor switches. In a Class D amplifier, the transistors switch between a high voltage, zero current cutoff state and the low voltage, high current saturation state. This produces very low steady-state losses. Unfortunately, significant power dissipation can occur during switching. For these reasons, Class D circuits have traditionally been used for low frequency, high power applications^3,4. In addition, Class D circuits have traditionally been used in pulse width modulation circuits, with low pass filters on the output to obtain a sine wave output. However, this paper shows that if the low pass filter is dispensed with, Class D switching circuits can be built for use as pulse amplifiers with switching times of a few nanoseconds.

The pulse amplifier presented in this paper was developed to provide a high speed voltage pulse source for reverse recovery measurements in diodes. Reverse recovery measurements are widely used as a tool in the study of diodes, since they can provide information about the diode structure and the carrier lifetimes inside the diode.

II. Amplifier Circuit

Figure 1 shows the amplifier circuit to be considered. It is based on a pulse amplifier chain presented by Krauss et al⁵ which was originally intended for a pulse width modulation circuit, with several modifications to allow for high speed operation. The level shifting diodes D2, D3 and D4 have been added to allow bipolar operation. The zener diodes D1 to D4 have been bypassed with large capacitors, which in effect means that the circuit is both dc-coupled through the zener diodes, and ac-coupled through the capacitors. The capacitors will present a low impedance to switching transients which will increase the transient base drive and decrease the transistor switching times, and the zener diodes provide for dc-coupling, which permits pulses of long duration to be amplified. Since the inputs of the Class D stages will have relatively low input impedances⁶, complementary emitter-follower stages have been added as voltage buffers. This reduces the output current required from the first Class D stage, which improves the switching speed. Also, the first Class D stage operates from lower power supply voltages than the second stage, to reduce the transistor power dissipation and switching time, since the full output voltage swing is not required to drive the second stage. Lastly, since a sine wave output is not desired, no tuned filter is present on the output.

FIG. 1. Schematic diagram of the pulse amplifier. The first Class D stage shapes a fast pulse to trigger the second Class D stage. Both stages are buffered by complementary emitter-followers to ease the drive requirements.

The power supply voltages for the circuit of Figure 1 are ±20V. The Zener diodes D5 to D8 drop these voltages to ±8.1V to power the first stage of the circuit. The breakdown voltage of the Zener diode D1 is 6.2V, and 10 V for D2. When the input voltage is zero, the voltage at the emitters of Q1 and Q2 (point C in Figure 1) will be approximately 0.7V, and there will be a negligible voltage drop across the base-emitter junction of Q4, so the diode D2 is reversed biased with approximately 8.8 V across it, which is less that its breakdown voltage. Thus almost no current flows in the diode, and transistor Q4 is in the cutoff state. However, D1 is in breakdown, since the base voltage of Q3 is approximately +8.1V-0.7V, and the voltage at point C is 0.7 V, yielding 6.7 V across D1 and R1, causing D1 to conduct. Thus Q3 is saturated, and the output voltage is +8.1V+VCEsat1 8 V.

When the input rises to the high level (+3.5V), D1 no longer has enough potential across it to sustain breakdown, and becomes non-conducting, forcing Q3 off. D2 is driven into breakdown, and conducts, turning Q4 on and driving it into saturation. The collector voltage falls to -8.1V+VCEsat2 -8 V. This yields a fast ±8V, inverted pulse at point D, which drives the second buffer and inverting Class D stages in a similar manner.

Figure 2 shows a typical output pulse for the circuit of Figure 1. It shows a ±20 V pulse into a 50 load, with rise and fall times of less than 3 ns. This is considerably better than what can be obtained with the fastest available op amps. By changing the Zener diodes D1 to D4, the output voltages can be easily changed to values other than +20 V and -20 V. In practice, it is found that the pulse repetition frequency should be kept below 1 MHz to ensure that the switching losses in the transistors do not become excessive and damage the transistors⁷.

FIG. 2. Typical output waveform for the circuit of Figure 1. Scale: 10 V/div, 10 ns/div.

III. Application to Reverse Transient Measurements

Figure 3 shows the circuit used for diode reverse recovery measurements presented here. An output waveform for the common 1N4148 fast switching diode is shown in Figure 4. At t < 30 ns, the diode is forward biased. At t = 30 ns, the voltage across the diode-resistor network is switched. For another 5 ns the diode appears as a low resistance due to the stored charge in the diode, and a large reverse current flows. After t = 35 ns, most of the stored charge has been removed, and the diode begins to accumulate a reverse voltage, and the diode current eventually falls to almost zero. This diode was chosen to illustrate the need for a high speed voltage pulse. Since the reverse transient is only several nanoseconds long, a fast pulse edge is required.

FIG. 3. Test circuit for reverse recovery transient measurements. The diode conducts a reverse current for a short time.

FIG. 4. Reverse recovery transient for a 1N4148 diode. The 1N4148 is a fast switching diode, as demonstrated by its very short reverse recovery transient. Scale: 10 V/div, 10 ns/div.

Figures 5 and 6 show reverse recovery transients for two other diodes, both 400 V fast-recovery power rectifiers. The diode used for Figure 5 is a TRW DSR3400X. The diode current shows a very "snappy" response, that is, the ratio of the constant reverse current time to the decaying reverse current time is very high. For most applications, a snappy transient is highly undesirable, since it can create large inductive voltage spikes. In contrast, the waveform for the Central Semiconductor 1N4936 shown in Figure 6 shows the classic textbook reverse recovery transient, with a constant current period followed by a very smooth fall in current.

FIG. 5. Reverse recovery transient for a TRW DSR3400X fast-recovery rectifier. Note the undesirable "snappy" response. Scale: 10 V/div, 10 ns/div.

FIG. 6. Reverse recovery transient for a Central Semiconductor 1N4936 fast-recovery rectifier. Note the classic textbook form of the reverse recovery transient. Scale: 10 V/div, 20 ns/div.

The nature of the reverse transient can be linked to the diode's doping profile. For instance, the snappy nature of the DSR3400X transient suggests that it is either has a diffused structure, or an epitaxial p+ p- n+ structure⁸. The smoother recovery of the 1N4936 suggest that it has an epitaxial p+ n- n+ structure⁸ or an epitaxial p+ n-- n- n+ structure^9,10. Figure 7 shows the approximate doping profile for the TRW DSR3400X, and Figure 8 shows the approximate doping profile for the 1N4936, with the junctions at x = 0. The profiles were obtained from high voltage C-V measurements¹¹, assuming that the junctions were one-sided (i.e. |N(x)| for x < 0). Figure 7 clearly shows that the DSR3400X is indeed a diffused structure, as the doping gradually varies from a very low level to a very high level. The 1N4936 clearly shows a p+ n-- n- n+ structure. That is, two lightly doped regions exist: the one closest to the junction is nearly intrinsic silicon with doping of less than 10¹² cm^-3 (n--), followed by a second layer of higher, but still quite light doping of around 10¹⁴ cm^-3 (n-). This structure is specifically designed to provide a smooth transient^9,10.

FIG. 7. Doping profile of the TRW DSR3400X fast-recovery rectifier. The doping profile is clearly diffused, as suggested by the snappy reverse recovery transient.

FIG. 8. Doping profile of the Central Semiconductor 1N4936 fast-recovery rectifier. The doping profile shows an active region consisting of an nearly intrinsic layer followed by a lightly doped layer. This modern design produces the smooth transient shown in Figure 6, rather than an abrupt transient like that shown in Figure 7.

References

¹ Comlinear Data Book, (Comlinear, Fort Collins, CO, 1993), p. 3-19.

² Hewlett-Packard Application Note 918 (Hewlett-Packard, Palo Alto, CA).

³ D. F. Page, W. D. Hindson, W. J. Chudobiak, Proc. IEEE 53, 423 (1965).

⁴ W. J. Chudobiak and D. F. Page, IEEE J. Solid-State Circ. SC-4, 25 (1969).

⁵ H. L. Krauus, C. W. Bostian, F. H. Raab, Solid State Radio Engineering (Wiley, New York, 1980), p. 467

⁶ H. R. Camenzind, IEEE Trans. Audio and Electroacoustics AU-14, 136 (1966).

⁷ K. K. Clarke and D. T. Hess, Communication Circuits: Analysis and Design (Addison-Wesley, Reading, MA, 1971), p. 418.

⁸ H. Benda and E. Spenke, Proc. IEEE 55, 1331 (1967).

⁹ E. D. Wolley and S. F. Bevacqua, IEEE Industrial Application Society Meeting Digest, (IEEE, Piscataway, NJ, 1981), p. 797.

¹⁰ B. J. Baliga, Modern Power Devices (Wiley-Interscience, New York, 1987), p. 416.

¹¹ M. J. Chudobiak, Rev. Sci. Instr. 66, 3703 (1995).